Reference signal channel estimation

ABSTRACT

Aspects of this disclosure relate to reference signal channel estimation. A wireless communication channel between two nodes can be estimated based on a received reference signal, such as a Sounding Reference Signal. Techniques are disclosed to improve performance of reference signal channel estimation and make channel estimates more robust in the presence of one or more of a variety of impairments. Frequency domain processing and/or time domain processing can be performed to reduce distortion in channel estimates.

BACKGROUND Technical Field

Embodiments of this disclosure relate to estimating a channel in acommunications system using a reference signal.

Description of Related Technology

In a wireless communication system, it can be desirable to have anaccurate estimation of a communication channel between a user equipmentand a base station. Certain communication standards include referencesignals that can be used to estimate a communication channel. Such areference signal can be used to estimate an uplink channel from a userequipment to a base station. In certain applications, a downlink channelfrom the base station to the user equipment to the base station can beestimated based on the uplink channel. In real world wirelesscommunication systems, a channel estimate can be degraded for a varietyof reasons.

SUMMARY OF CERTAIN INVENTIVE ASPECTS

The innovations described in the claims each have several aspects, nosingle one of which is solely responsible for its desirable attributes.Without limiting the scope of the claims, some prominent features ofthis disclosure will now be briefly described.

One aspect of this disclosure is a method of reference signal channelestimation. The method comprises receiving a reference signal forchannel estimation; de-covering the reference signal in a frequencydomain to generate a de-covered reference signal; after the de-covering,frequency domain processing the de-covered reference signal to causedistortion of a direct current offset in the de-covered reference signalto be reduced; after the frequency domain processing, time domainprocessing to cause a noise floor associated with the de-coveredreference signal to be reduced; and generating a channel estimate basedon the frequency domain processing and the time domain processing,wherein the channel estimate is associated with a communication channelbetween a first node and a second node.

The frequency domain processing can comprise generating an estimatedtone for a tone of the de-covered frequency domain reference signalbased on at least two other tones of the de-covered frequency domainreference signal. The frequency domain processing can comprise replacingthe tone with the estimated tone to cause distortion associated with thedirect current offset to be reduced. The frequency domain processing cancomprise modifying the tone based on the estimated tone to causedistortion associated with the direct current offset to be reduced.

The frequency domain processing can comprise pulse shaping thede-covered frequency domain reference signal to cause distortionassociated with the direct current offset to be reduced.

The time domain processing can comprise estimating noise power for asub-set of time domain taps corresponding to sub-carriers betweenchannel impulse responses, and performing a per-tap scaling on at leasta portion of the time domain taps based on the estimating. The per-tapscaling can involve minimum mean squared error scaling. The per-tapscaling can involve thresholding.

The method can further comprise pulse shaping the reference signal priorto the de-covering.

The method can further comprise rotating the reference signal based onan indicator of a frequency offset prior to the de-covering.

The time domain processing can comprise moving a spur outside of timedomain windows for cyclic shifts of the reference signal. The method canfurther comprise pulse shaping the reference signal prior to thede-covering.

The reference signal can be an uplink Sounding Reference Signal.

The first node can be a user equipment and the second node can be anetwork node. The first node can be a user equipment and the second nodecan include a remote radio unit. The first node can be a user equipmentand the second node can include a base station integrated with anantenna front-end. The first node can include a first remote radio unitand the second node can include a second remote radio unit. The firstnode can be a first user equipment and the second node can be a seconduser equipment.

Another aspect of this disclosure is a system for channel estimation.The system comprises a frequency domain processing circuit, a timedomain processing circuit, and a channel estimation circuit. Thefrequency domain processing circuit is configured to generate ade-covered frequency domain reference signal and to process thede-covered frequency domain reference signal so as to cause distortionassociated with a direct current offset to be reduced. The time domainprocessing circuit has an input coupled to an output of the frequencydomain processing circuit. The time domain processing circuit isconfigured to suppress time domain channel impulse response leakage. Thechannel estimation circuit is configured to generate a channel estimatebased on an output of the time domain processing circuit, in which thechannel estimate is associated with a wireless communication channelbetween a first node and a second node.

The frequency domain processing circuit can be configured to generate anestimated tone for a tone of the de-covered frequency domain referencesignal based on at least two other tones of the de-covered frequencydomain reference signal, and to replace the tone with the estimated toneto cause distortion associated with the direct current offset to bereduced.

The frequency domain processing circuit can be configured to performpulse shaping on the de-covered frequency domain reference signal tocause distortion associated with the direct current offset to bereduced.

The time domain processing circuit can comprise: a filter comprising aplurality of taps; a noise power estimation circuit configured toestimate noise power for a sub-set of the taps of the filtercorresponding to sub-carriers between channel impulse responses; and afilter tap scaling circuit configured to perform a per-tap scaling on atleast a portion of the taps of the filter based on the estimated noisepower.

The time domain processing circuit can be configured to move a spuroutside of time domain windows for cyclic shifts of the referencesignal.

The system can further comprise a second time domain processing circuitconfigured to pulse shape a reference signal, the second time domainprocessing circuit having an output coupled to an input of the frequencydomain processing circuit.

The system can further comprise a second time domain processing circuitconfigured to rotate a reference signal based on an indicator of afrequency domain offset to thereby reduce the frequency domain offset,in which the second time domain processing circuit has an output coupledto an input of the frequency domain processing circuit.

The system can further comprise a second frequency domain processingcircuit configured to perform per-cyclic shift frequency domainprocessing, wherein the second frequency domain processing circuit iscoupled between the time domain processing circuit and the channelestimation circuit.

The first node can be a user equipment and the second node can be anetwork node. The first node can be a user equipment and the second nodecan include a remote radio unit. The first node can be a user equipmentand the second node can include a base station integrated with anantenna front-end.

The de-covered frequency domain reference signal can be a de-covereduplink Sounding Reference Signal.

Another aspect of this disclosure is a system for channel estimationthat comprises: means for processing a de-covered frequency domainreference signal so as to cause distortion associated with a directcurrent offset to be reduced; means for suppressing time domain channelimpulse response leakage, the means for suppressing having an inputcoupled to an output of the means for processing; and a channelestimation circuit configured to generate a channel estimate based on anoutput of the means for suppressing, wherein the channel estimate isassociated with a wireless communication channel between a first nodeand a second node.

For purposes of summarizing the disclosure, certain aspects, advantagesand novel features of the innovations have been described herein. It isto be understood that not necessarily all such advantages may beachieved in accordance with any particular embodiment. Thus, theinnovations may be embodied or carried out in a manner that achieves oroptimizes one advantage or group of advantages as taught herein withoutnecessarily achieving other advantages as may be taught or suggestedherein.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of this disclosure will now be described, by way ofnon-limiting example, with reference to the accompanying drawings.

FIG. 1 is a schematic block diagram associated with a reference signalmodel.

FIG. 2 is a schematic block diagram of hardware of a user equipment ofFIG. 1 that is arranged to generate a reference signal.

FIG. 3 is a schematic block diagram of processing circuitry arranged togenerate a channel estimate.

FIG. 4A is a schematic block diagram of a first part of processingcircuitry arranged to generate a channel estimate.

FIG. 4B is a schematic block diagram of a second part of processingcircuitry arranged to generate a channel estimate.

FIG. 5 is a diagram that illustrates time domain channel impulseresponse separation for a reference signal with 8 cyclic shifts.

FIGS. 6A to 6E are block diagrams of parts of the processing circuitryof FIGS. 4A to 4B that can improve channel estimation according tocertain embodiments.

FIG. 7A is a graph that illustrates a magnitude of a de-covered SoundingReference Signal from two cyclic shifts with some distortion in directcurrent and low frequency tones.

FIG. 7B is a graph illustrating distorted direct current and lowfrequency tones in a time domain.

FIG. 8A is a graph illustrating a channel frequency response in afrequency domain.

FIG. 8B is a graph illustrating power leaking in a time domainassociated with the channel frequency response of FIG. 8A.

FIG. 9 is a graph that illustrates frequency domain pulses with a timingoffset, a frequency offset, and no timing or frequency offset.

FIG. 10 shows an example of interpolation for reference signal channelestimation.

FIG. 11 shows an example of extrapolation for reference signal channelestimation.

FIG. 12A shows a received frequency domain SRS after an FFT is performedwith distorted low-frequency tones.

FIG. 12B shows an SRS after applying quadratic interpolation to 16 lowfrequency tones.

FIG. 12C shows an SRS after applying quadratic extrapolation to 16 edgetones on each side.

FIG. 13A shows window functions for a rectangle pulse and a raisedcosine pulse.

FIG. 13B shows time domain tap powers relative to a center tap for therectangle and raised cosine pulses respectively.

FIG. 14A is a plot of an SRS with extrapolated edge tones beforefrequency domain pulse shaping.

FIG. 14B is a plot of a frequency domain pulse-shaping function.

FIG. 14C is a plot of the SRS that includes extrapolated edge tonesafter frequency domain pulse shaping is applied.

FIG. 15 is a diagram that illustrates time domain channel impulseresponses for cyclic shifts after an inverse FFT has been performed.

FIG. 16 is a flow diagram of an example method of time domain channelimpulse response scaling based on a power delay profile.

FIG. 17 is a schematic block diagram of a time domain processing circuitthat can perform time domain channel impulse response scaling based on apower delay profile.

FIG. 18 is a schematic block diagram of processing circuitry with timedomain pulse shaping.

FIG. 19A is a graph of a rectangle pulse and a raised cosine pulse inthe time domain.

FIG. 19B is a graph of a sinc pulse and a raised cosine pulse in thefrequency domain.

FIG. 20 is a schematic block diagram of processing circuitry withfrequency rotation to compensate for a frequency offset.

FIG. 21 is a flow diagram of a method of detecting cyclic shifts of areference signal in which spurs are moved into an unused time space.

FIG. 22 illustrates the locations of the first harmonic spurs for fourcyclic shifts.

FIG. 23 is a diagram illustrating an example multiple-inputmultiple-output (MIMO) network environment in which channel estimationbased on a reference signal can be performed.

DETAILED DESCRIPTION OF CERTAIN EMBODIMENTS

The following description of certain embodiments presents variousdescriptions of specific embodiments. However, the innovations describedherein can be embodied in a multitude of different ways, for example, asdefined and covered by the claims. In this description, reference ismade to the drawings where like reference numerals can indicateidentical or functionally similar elements. It will be understood thatelements illustrated in the figures are not necessarily drawn to scale.Moreover, it will be understood that certain embodiments can includemore elements than illustrated in a drawing and/or a subset of theelements illustrated in a drawing. Further, some embodiments canincorporate any suitable combination of features from two or moredrawings. The headings provided herein are for convenience only and donot necessarily affect the scope or meaning of the claims.

In Long Term Evolution (LTE) and New Radio (NR) systems, a SoundingReference Signal (SRS) can be transmitted from a user equipment (UE) toa Node-B for estimating of an uplink channel. More generally, an SRS canbe transmitted from an antenna of a first node to an antenna of a secondnode. For time-division duplex (TDD) systems, the SRS channel estimate(CE) can be used to estimate a downlink channel quality based on channelreciprocity. In multiple-input multiple-output (MIMO) systems based oncodebooks and/or regularized zero-forcing (RZF) precoding, SRS CE can beused in selecting a precoding matrix. In such instances, it is desirablefor the accuracy of SRS CE to be high, especially in RZF precodingapplications. SRS CE can be used for calibration.

There are a variety of impairments in a real word system that maydegrade the performance of SRS CE. Such impairments include withoutlimitation one or more of frequency offset, timing offset, time domainchannel impulse response (CIR) leakage, or distortion of relatively lowfrequency tones. In addition, the output SRS CE of edge tones may beinaccurate due to a discontinuity of the SRS in the frequency domain. Inthis disclosure, techniques to improve the performance of SRS CE areprovided to make channel estimates more robust in the presence of one ormore impairments.

Sounding Reference Signal

In LTE and fifth generation (5G) NR, the construction of an SRStypically includes SRS sequence generation and the mapping to physicalresources. SRS can be generated according to a relevant communicationstandard. SRS can be an uplink SRS in any suitable embodiment disclosedherein.

In LTE and/or NR uplink transmissions, there can be dedicated orthogonalfrequency domain multiplexing (OFDM) symbols for SRS transmission toscheduled UEs. In each symbol, mSRS resource blocks for SRS can beshared by a group of UEs. All sub-carriers in the mSRS resource blocksfor SRS can be divided into K_(TC) orthogonal combs. An orthogonal combcan be a group of signals allocated at a given time that are orthogonalin the frequency domain. The number of resource elements M_(ZC) in eachcomb can be in accordance with Equation 1, where M_(ZC) can represent alength of an SRS sequence.

M _(ZC) =m _(SRS) N _(SC) ^(RB) /K _(TC)  (Eq.1)

In Equation 1, N_(SC) ^(RB) is the number of sub-carriers in eachresource block. N_(SC) ^(RB) can be 12, for example. Within each comb,M_(ZC) resource elements are shared by up to n_(SRS) ^(CS,max) SRSsthrough different cyclic shifts. The transmission comb number K_(TC), isspecified by the base station and can take the value of 2 or 4. Themaximum number of cyclic shifts n_(SRS) ^(CS,max) that can be supportedby one comb can be a function of K_(TC) as indicated by Equation 2.

$\begin{matrix}{n_{SRS}^{{cs},\max} = \left\{ \begin{matrix}{8,} & {K_{TC} = 2} \\{12,} & {K_{TC} = 4}\end{matrix} \right.} & \left( {{Eq}.\mspace{14mu} 2} \right)\end{matrix}$

For the UE to transmit an SRS over the mSRS resource blocks, the UEshould generate an SRS sequence of length M_(ZC). The first step for theUE can be to obtain a base sequence r_(u) ,(n) of length M_(ZC), where ndenotes the index of the allocated tones. For a given length M_(ZC),there can be 30 or 60 base sequences r_(u,v) (n) that are divided intogroups, where u ∈ {0, 1, . . . , 29} is the group number. There can beone or two sequences in one group, and the base sequence index within agroup can be v={0} or v={0, 1}, respectively.

For M_(ZC)≥36, r_(u,v) (n) can be obtained from a Zadoff-Chu (ZC)sequence by Equation 3.

r _(u,v)(n)=x _(q)(n mod N _(ZC))  (Eq. 3)

In Equation 3, N_(ZC) can be the largest prime number such thatN_(ZC)<M_(ZC) and x_(q)(m) is a ZC sequence of length N_(ZC) that isobtained from Equations 4-1 to 4-3.

$\left\{ {\quad\begin{matrix}{{x_{q}(m)} = {e^{{- j}\frac{\pi\;{{qm}{({m + 1})}}}{N_{ZC}}}\mspace{461mu}\left( {{{Eq}.\mspace{14mu} 4} - 1} \right)}} \\{q = {\left\lfloor {\overset{\_}{q} + {1/2}} \right\rfloor + {{v \cdot \left( {- 1} \right)^{\lfloor{2\overset{\_}{q}}\rfloor}}\mspace{385mu}\left( {{{Eq}.\mspace{14mu} 4} - 2} \right)}}} \\{\overset{\_}{q} = {{N_{ZC} \cdot {\left( {u + 1} \right)/31}}\mspace{455mu}\left( {{{Eq}.\mspace{14mu} 4} - 3} \right)}}\end{matrix}} \right.$

The r_(u,v) (n) for MZC<36 can be generated according to a 3rdGeneration Partnership Project (3GPP) standard.

Group and sequence hopping for SRS can be configured. In group orsequence hopping, a base sequence is selected by varying u and v foreach SRS symbol. In many applications, group or sequence hopping can bedisabled and the calculation of u and v is simplified as shown inEquations 5-1 and 5-2, respectively.

$\left\{ {\quad\begin{matrix}{u = {n_{ID}^{SRS}\mspace{14mu}{mod}\mspace{14mu} 30\mspace{481mu}\left( {{{Eq}.\mspace{14mu} 5} - 1} \right)}} \\{v = {0\mspace{616mu}\left( {{{Eq}.\mspace{14mu} 5} - 2} \right)}}\end{matrix}} \right.$

In Equation 5-1, n_(ID) ^(SRS) is given by the higher layer. In thefollowing, it is assumed that group or sequence hopping is disabled andhence the base sequence is denoted as r(n) with the subscripts u and vbeing omitted.

After the base sequence r(n) is obtained in each UE, the SRS sequencer^((c))(n) can be generated by applying a cyclic shift in frequencydomain according to Equation 6.

$\begin{matrix}\begin{matrix}{{r^{(c)}(n)} = e^{{J\; 2\;{\pi \cdot {({c/n_{SRS}^{{cs},\max}})}}{n \cdot {\overset{\_}{r}{(n)}}}},}} & {{{for}\mspace{14mu} 0} \leq n < {M_{ZC}\mspace{14mu}{and}\mspace{14mu} 0} \leq c < n_{SRS}^{{cs},\max}}\end{matrix} & \left( {{Eq}.\mspace{14mu} 6} \right)\end{matrix}$

In Equation 6, c is the cyclic-shift index. Then, r^((c))(n) are mappedto an allocated comb, converted to the time domain signal through aninverse Fast Fourier Transform (IFFT). The time domain signal can betransmitted over channels with channel impulse response (CIR) h_(c)(t),0≤c<n_(SRS) ^(CS,max), to a specific antenna. An overall block diagramfor the SRS model is shown in FIG. 1.

FIG. 1 is a schematic block diagram associated with a reference signalmodel. In FIG. 1, a plurality of UEs 10A, 10B, 10N wirelessly transmitSRSs with different cyclic shifts over respective channels 12A, 12B, 12Nto a Node B 14. The different cyclic shifts can identify which deviceand/or antenna is wirelessly transmitting an SRS. At a summing point 13,the SRSs can be summed in the reference signal model. The Node-B 14 caninclude a radio frequency (RF) downconverting block 16 and a samplingand cyclic prefix (CP) removal block 18. The RF downconverting block 16can downconvert a received RF signal. This downconversion can be tobaseband, for example. The sampling and CP removal block 18 can samplethe downconverted RF reference signal. The sampling and CP removal block18 can remove a cyclic prefix. The Node-B 14 can generate time domainreceive samples from SRSs received from the UEs 10A, 10B, 10N. The NodeB 14 can be an evolved Node B (eNode B), a next generation Node B (gNodeB) or replaced and/or implemented together with any suitable basestation or network system.

With the maximum number of cyclic shifts being 8, there can be 8 UEs and8 channels in the reference signal model. Similarly, with the maximumnumber of cyclic shifts being 12, there can be 12 UEs and 12 channels inthe reference signal model. Any other suitable number of maximum cyclicshifts and corresponding channels can be implemented in accordance withany suitable principles and advantages disclosed herein.

Although UEs are illustrated in FIG. 1, the principles and advantagesdisclosed herein can be applied to channel estimation between anysuitable nodes arranged to wirelessly communicate with each other. Forexample, SRS channel estimation can be used to estimate a channelbetween an UE and a network node, for example, as shown in FIG. 1. Thenetwork node can be a base station integrated with an antenna front-end.The network node can include a remote radio unit (RRU) and a base bandunit (BBU). As another example, SRS channel estimation can be used toestimate a channel between two network nodes (e.g., between 2 nodes thatinclude RRUs). As one more example, SRS channel estimation can be usedto estimate a channel between two UEs. SRS channel estimation inaccordance with any suitable principles and advantages disclosed hereincan be applied to channel estimation between any suitable nodes arrangedto wirelessly communicate information.

Although embodiments disclosed herein may be described with reference tothe SRS for illustrative purposes, any suitable principles andadvantages disclosed herein can be applied to channel estimation usingany suitable reference signal and/or any suitable pilot signal.

FIG. 2 is a schematic block diagram of hardware of a UE of FIG. 1 thatis arranged to generate a reference signal. Each UE 10A to 10N of FIG. 1can include the hardware shown in FIG. 2. The functionality describedwith reference to the blocks of FIG. 2 can be implemented by anysuitable physical hardware. The functionality described with referenceto FIG. 2 can be implemented in any suitable node arranged to transmitan SRS and/or other reference signal.

As illustrated in FIG. 2, reference signal generation circuitry caninclude a base sequence generator 21, a phase ramping block 22, aresource element mapping block 23, an IFFT block 24, and adigital-to-analog conversion and RF modulation block 25. The basesequence generator 21 can generate a base sequence, for example, asdescribed above. As one example, the base sequence can be generatedusing Equation 3. The phase ramping block 22 can apply a cyclic shift tothe base sequence generated by the base sequence generator 21. Thecyclic shift can be applied in the frequency domain. Each node (e.g.,each UE 10A to 10N in FIG. 1) can have a different cyclic shift appliedby a respective phase ramping block 22. The resource element mappingblock 23 can map the SRS sequence from the phase ramping block 22 to anallocated comb. Then the IFFT block 24 can convert the frequency domaincyclic shifted signal to the time domain by an inverse Fouriertransform. A cyclic prefix can be added by the IFFT block 24. Thedigital-to-analog conversion and RF modulation block 25 can convert theoutput of the IFFT block 24 to an analog signal and modulate the analogsignal to a radio frequency. An RF signal provided by thedigital-to-analog conversion and RF modulation block 25 can bewirelessly transmitted over a communication channel to a Node B and/orother suitable hardware for channel estimation.

Reference Signal Channel Estimation

Processing circuitry can receive a reference signal, such as an SRS, andgenerate a channel estimate based on the received reference signal. Thechannel estimate can be for a relatively wide band SRS. Example signalprocessing will be described with reference to FIGS. 3 to 6E. The signalprocessing circuitry can perform improved reference signal channelestimation in accordance with any suitable principles and advantagesdisclosed herein.

FIG. 3 is a schematic block diagram of processing circuitry 30 arrangedto generate a channel estimate. The processing circuitry 30 can receivetime domain reference signal samples and generate a channel estimatebased on the reference signal. The time domain samples can be received,for example, from a sampling and CP removal block 18 of FIG. 1. Asillustrated in FIG. 3, the processing circuitry 30 includes a FastFourier Transform (FFT) block 31, a first frequency domain processingcircuit 32, an inverse Fast Fourier Transform (IFFT) block 33, a timedomain processing circuit 34, a second FFT block 35, a second frequencydomain processing circuit 36, and a channel estimation circuit 37.

The received reference signal can be an SRS. The received SRS can beconverted into frequency domain by the FFT block 31. The frequencydomain processing circuit 32 has an input coupled to an output of theFFT block 31. The first frequency domain processing circuit 32 canextract SRS symbols of a comb for de-covering. The first frequencydomain processing circuit can de-cover the reference signal using a basesequence. The de-covered frequency domain reference signal can contain asum of channel frequency responses with different phase ramping.Additional frequency domain processing can be performed by the firstfrequency domain processing circuit 32 after de-covering the referencesignal to improve SRS channel estimate performance. The first frequencydomain processing circuit 32 can perform frequency domain processing tocause distortion of a direct current offset and/or a low frequencyoffset to be reduced. Examples of such processing will be discussedbelow.

The IFFT block 33 can convert the de-covered frequency domain signalfrom the first frequency domain processing circuit 32 to the timedomain. An output signal from the IFFT block 33 can be provided to thetime domain processing circuit 34. The time domain processing circuit 34can perform time domain processing to improve SRS channel estimateperformance. The time domain processing circuit 34 can perform timedomain processing to cause a noise floor to be reduced. Examples of suchprocessing will be discussed below. The time domain processing circuit34 can separate channel impulse responses. Separating channel impulseresponses can be performed in accordance with any suitable principlesand advantages discussed with reference to FIG. 6.

The second FFT block 35 can convert an output signal from the timedomain processing circuit 34 to the frequency domain. The secondfrequency domain processing circuit 36 can perform per cyclic shiftfrequency domain processing. The second frequency domain processingcircuit 36 can output a channel frequency response for each cyclicshift.

The channel estimation circuit 37 can generate a channel estimate basedon an output of the second frequency domain processing circuit 36. Thechannel estimate is based on the reference signal received by theprocessing circuitry 30. The channel estimate is associated with awireless communication channel between a first node and a second node.As one example, the first node can be a UE 10A of FIG. 1 and the secondnode can be the Node B 14 of FIG. 1. Processing techniques disclosedherein can improve the channel estimate generated by the channelestimation circuit 37. This can be advantageous in a variety ofapplications, such as for selecting a precoding matrix in TDD MIMOsystems.

More details regarding embodiments of the processing circuitry 30 areprovided in FIGS. 4A and 4B and the corresponding description. FIG. 4Ais a schematic block diagram of a first part of processing circuitry 40for generating a channel estimate. FIG. 4B is a schematic block diagramof a second part of the processing circuitry 40.

The processing circuitry 40 receives a reference signal, such as an SRS.The first FFT block 31 converts the received reference signal into thefrequency domain. A reference signal extraction circuit 42 can extractall resource elements of one comb for de-covering. This can involveextracting SRS symbols of a current comb. The summed SRSs in thefrequency domain can be represented by Equation 7. In Equation 7,H_(c)(n)=(h_(c)(kT_(s))) is the channel frequency response (CFR) ofcyclic-shift c, and v(n) denotes the additive white Gaussian noise(AWGN).

$\begin{matrix}{{{y(n)} = {{\sum\limits_{c = 0}^{n_{SRS}^{{cs},\max}}{e^{{j2}\;{\pi \cdot {({c/n_{SRS}^{{cs},\max}})} \cdot n}} \cdot {\overset{\_}{r}(n)} \cdot {H_{c}(n)}}} + {v(n)}}},{0 \leq n < M_{ZC}}} & \left( {{Eq}.\mspace{14mu} 7} \right)\end{matrix}$

A base sequence generator 43 generates a base sequence for de-covering.The base sequence can be generated in accordance with any suitableprinciples and advantages disclosed herein. A mixer 44 or any othersuitable circuit can be used to de-cover the reference signal using thebase sequence from the base sequence generator 43.

After being de-covered by the base sequence r(n), the de-coveredfrequency domain signal can be represented by Equation 8. Equation 8shows that the de-covered frequency domain signal contains a sum ofchannel frequency responses with different linear phase ramping.

$\begin{matrix}{{{y^{\prime}(n)} = {{{y(n)} \cdot {{\overset{\_}{r}}^{*}(n)}} = {{\sum\limits_{c = 0}^{n_{SRS}^{{cs},\max} - 1}{e^{{j2}\;{{\pi(\frac{c}{n_{SRS}^{{cs},\max}})} \cdot n}} \cdot {H_{c}(n)}}} + {v(n)}}}},{0 \leq n < M_{ZC}}} & \left( {{Eq}.\mspace{14mu} 8} \right)\end{matrix}$

A frequency domain processing circuit 45 can perform frequency domainprocessing on the de-covered signal to improve reference signal channelestimate performance.

Denote h(k) as a time-domain signal of length N and define h(S)(k) asthe sequence by cyclic-shifting h(k) to the right by S, i.e.,h^((s))(k)=h(MOD(k+S, N)), for 0≤k<N. The de-covered signal y′(n) can beconverted into time domain through an IFFT by the IFFT block 33. In thetime domain, the signal becomes a combination of multiple channelimpulse responses with different cyclic shifts, which can be representedby Equation 9. In Equation 9, N_(IFFT) represents the IFFT size percomb.

$\begin{matrix}{{{s(k)} = {{{IFFT}\left( {y^{\prime}(n)} \right)} = {{\sum\limits_{c = 0}^{n_{SRS}^{{cs},\max} - 1}{{h_{c}}^{({\frac{c}{n_{SRS}^{{cs},\max}}N_{IFFT}})}(k)}} + {w(k)}}}},{0 \leq k < N_{IFFT}}} & \left( {{Eq}.\mspace{14mu} 9} \right)\end{matrix}$

After the IFFT, time domain processing can be performed by the timedomain processing and channel impulse response separation block 48 ofFIG. 4B to improve the SRS CE performance. Assuming that the maximumdelay spread of any h_(c) is less than N_(IFFT)/n_(SRS) ^(cs,max), thetime-domain channel for different cyclic shifts can be separatedrelatively easily by the time domain processing and channel impulseresponse separation block 48. The second FFT block 35 can includesub-FFT blocks 35A to 35N to transform individual channel impulseresponses to the frequency domain. Per cyclic shift frequency domainprocessing circuits 50A to 50N can obtain the channel frequencyresponses for each cyclic shift. A channel estimation circuit, such asthe channel estimation circuit 37 of FIG. 3, can generate channelestimates based on the channel frequency responses for the cyclicshifts.

Separating channel impulse responses will now be discussed. Suchfunctionality can be performed, for example, by the time domainprocessing and channel impulse response separation block 48 of FIG. 4B.To separate channel impulse responses, a window can be defined for eachof the cyclic shifts. For a specific cyclic shift, only taps fallinginto the window for the specific cyclic shift are preserved and allother taps can be set to zero. Then, the non-zero taps arecyclic-shifted, and in the frequency domain, the linear phase rampingcan be removed.

FIG. 5 is a diagram that illustrates time domain channel impulseresponse separation for a reference signal with 8 cyclic shifts. Theexample shown in FIG. 5 illustrates how to separate the channel impulseresponse for a specific shift (i.e., c=7) from other cyclic shifts. Thetop most portion of FIG. 5 shows a time domain impulse response. Thereare time domain impulse responses for 8 different cyclic shifts shown inFIG. 5. There are different windows defined for each of the cyclicshifts. Windowing can be performed by preserving a specific channelimpulse response within a specific window (i.e., the channel impulseresponse for cyclic shift c=7 in FIG. 5) and removing the other channelimpulse responses. The impulse response for the specific channel can beshifted to remove the cyclic shift. This can remove linear phase rampingin the frequency domain.

Properties of ZC sequences in SRS will now be discussed. If a ZCsequence x_(q) (m) defined in Equations 4-1 to 4-3 is de-covered by ashifted version of itself, denoted as x_(q) (m−s), s=0, ±1, ±2, ±3, . .. , the de-covered signal can be represented by Equation 10. In Equation10,

$\varphi = {\frac{nq}{\text{?}}\left( {s^{2} - s} \right)}$?indicates text missing or illegible when filed

can be a constant phase.

$\begin{matrix}{{{x_{q}(m)} \cdot {x_{q}^{*}\left( {m - s} \right)}} = {e^{{- j}\frac{2{nsqm}}{N_{ZC}}} \cdot e^{j\varphi}}} & \left( {{Eq}.\mspace{14mu} 10} \right)\end{matrix}$

Equation 10 reveals that x_(q) (m)·x_(q)*(m−s) is a sequence of linearphase ramping, with the slope

$\mspace{20mu}\frac{2\pi\;{sq}}{\text{?}}$?indicates text missing or illegible when filed

being a function of s and q. If the frequency hopping and sequencehopping are disabled, then q becomes dependent on cell ID.

The IFFT of x_(q) (m)·x_(q)*(m−s) represents an impulse in time domainand the delay of the impulse is related to s and q.

Frequency domain processing techniques can be applied to improvereference signal channel estimation. Such frequency domain processingcan reduce direct current and/or other low frequency offsets.Alternatively or additionally, frequency domain processing circuits cancompensate for any other suitable impairment to accurate referencesignal channel estimation. The frequency domain processing circuit 32 ofFIG. 3 and/or the frequency domain processing circuit 45 of FIG. 4A canimplement frequency domain processing to improve reference signalchannel estimation. FIGS. 6A, 6B, and 6C illustrate example blockdiagrams of frequency domain processing circuits that can be implementedby the frequency domain processing circuit 32 and/or the frequencydomain processing circuit 45. The frequency domain processing circuit 36of FIG. 3 and/or any of the frequency domain processing circuits 50A to50N of FIG. 4A can implement frequency domain processing to improvereference signal channel estimation. FIG. 6D illustrates example blockdiagram of a frequency domain processing circuit that can be implementedby the frequency domain processing circuit 36 of FIG. 3 and/or any ofthe frequency domain processing circuits 50A to 50N of FIG. 4B.

FIG. 6A is a block diagram of a frequency domain processing circuit 62according to an embodiment. The frequency domain processing circuit 62includes a tone estimation circuit 63 arranged to generate an estimatedtone for a tone of a de-covered frequency domain reference signal basedon at least two other tones of the de-covered frequency domain referencesignal. The tone estimation circuit 63 is also arranged to replace thetone with the estimated tone to cause distortion associated with thedirect current offset to be reduced. The tone estimation circuit 63 iscan alternatively or additionally modify the tone based on the estimatedtone to cause distortion associated with the direct current offset to bereduced. Tone estimation can involve interpolation. Tone estimation caninvolve extrapolation. Tone estimation can be based on poly-phasedecomposition. Tone estimation can be based on least squares estimation.Tone estimation can be based on any other suitable technique. Thefrequency domain processing circuit 62 can perform any other suitablefrequency domain processing for channel estimation, such as any othersuitable features discussed with reference to the frequency domainprocessing circuit 32 of FIG. 3 and/or the frequency domain processingcircuit 45 of FIG. 4A.

FIG. 6B is a block diagram of a frequency domain processing circuit 64according to an embodiment. The frequency domain processing circuit 64includes a channel frequency response pulse shaping circuit 65 arrangedto pulse shape a de-covered frequency domain reference signal. Pulseshaping can cause distortion associated with a direct current offset tobe reduced. The pulse shaping can involve a raised-cosine pulse and/orany other suitable pulse. The frequency domain processing circuit 64 canperform any other suitable frequency domain processing for channelestimation, such as any other suitable features discussed with referenceto the frequency domain processing circuit 32 of FIG. 3 and/or thefrequency domain processing circuit 45 of FIG. 4A.

FIG. 6C is a block diagram of a frequency domain processing circuit 66according to an embodiment. The frequency domain processing circuit 66includes the tone estimation circuit 63 and the channel frequencyresponse pulse shaping circuit 65. FIG. 6C illustrates that toneestimation can be implemented together with channel frequency responsepulse shaping. The frequency domain processing circuit 65 can performany other suitable frequency domain processing for channel estimation,such as any other suitable features discussed with reference to thefrequency domain processing circuit 32 of FIG. 3 and/or the frequencydomain processing circuit 45 of FIG. 4A.

FIG. 6D is a block diagram of a frequency domain processing circuit 67according to an embodiment. The frequency domain processing circuit 67includes a scaling circuit 68. The scaling circuit 68 can be implementedtogether with the channel frequency response pulse shaping circuit 65 ofFIGS. 6B and/or 6C. The scaling circuit 68 can scale the channelfrequency response of edge tones to compensate for the impact offrequency domain pulse shaping. The frequency domain processing circuit67 can perform any other frequency domain processing, such as any othersuitable features discussed with reference to the frequency domainprocessing circuit 36 of FIG. 3 and/or any of the frequency domainprocessing circuits 50A to 50N of FIG. 4B.

Time domain processing techniques can be applied to improve referencesignal channel estimation. In certain instances, time domain processingcan be implemented together with one or more frequency domain processingtechniques disclosed herein to improve reference signal channelestimation.

FIG. 6E is a block diagram of a time domain processing circuit 69according to an embodiment. The time domain processing circuit 69 isarranged to perform time domain processing and channel impulse responseseparation. The time domain processing circuit 69 includes a channelimpulse response scaling circuit 70. The channel impulse responsescaling circuit 70 is arranged to scale a channel impulse response basedon a power delay profile. The scaling can be applied per tap of afilter. The time domain processing circuit 69 can perform any other timedomain processing, such as any other suitable features discussed withreference to the time domain processing circuit 34 of FIG. 3 and/or thetime domain processing and channel impulse response separation block 48of FIG. 4B.

A time domain processing circuit can move a spur outside of time domainwindows for cyclic shifts of the reference signal in certainembodiments. For instance, harmonic spurs can be moved to unused timedomain space that does not impact channel estimation. Spurs can be movedto time domain indices below time domain windows for the cyclic shiftsand/or to time domain indices between time domain windows for cyclicshifts.

In certain applications, time domain pulse shaping can be applied beforea reference signal is translated into the frequency domain. Time domainpulse shaping can reduce frequency offset and/or timing offset.

According to some applications, frequency rotation can be applied beforea reference signal is translated into the frequency domain. This canreduce frequency offset.

Impairments on Reference Signal Channel Estimation

In a real world communication system, there can be a variety ofimpairments that may affect the performance of reference signal channelestimation. Such impairments can include one or more of distortion oflow frequency tones, time domain channel impulse response leakage,frequency offset, or timing offset. There can be a variety of factorsthat can result in inaccurate channel estimation. Identifyingimpairments that affect performance of reference signal channelestimation can be challenging. Impairments discussed herein wereidentified through analysis of SRS channel estimation data.

Distortion of low frequency tones will now be discussed. For an RF frontend with a zero intermediate frequency (ZIF) transceiver, direct current(DC) offset may be present in a receive baseband signal due to localoscillator (LO) leakage. To mitigate the DC offset component, a notchfilter can be applied to a demodulated signal before generating basebandsamples for digital signal processing. However, the notch filter may notwork perfectly and/or may distort the DC tone and one or more adjacenttones, especially in TDD systems where uplink symbols are not continuousin time.

If a DC tone and adjacent tones to the DC tone are distorted, colorednoise can be introduced with locations at DC and low frequency tones inthe frequency domain. The colored noise tones can elevate a noise floorin the time domain and introduce inter-cyclic shift interference in SRSCE.

FIG. 7A is a graph that illustrates a magnitude of a de-covered SoundingReference Signal from two cyclic shifts with some distortion in DC andlow frequency tones. The distorted tones in FIG. 7A are located at andnear DC.

FIG. 7B is a graph illustrating distorted DC and low frequency tones ina time domain. Sample power is plotted versus time index in FIG. 7B. Thedistorted tones contribute to a hump in the time domain in the curve ofFIG. 7B. The distorted tones are located between channel impulseresponses corresponding to different cyclic shifts (i.e., c=0 and c=1 inFIG. 7B). These distorted tones can raise the noise floor.

Time domain channel impulse response leakage will now be discussed. Whena frequency domain channel frequency response is transformed into thetime domain through an IFFT, the channel impulse response pulse may havea significant sidelobe leakage that goes into time domain windows forone or more other cyclic shifts.

FIG. 8A is a graph illustrating a channel frequency response in afrequency domain. The channel frequency response for the cyclic shiftc=0 is shown in the frequency domain. In FIG. 8A, the channel isrelatively flat in the frequency domain.

FIG. 8B is a graph illustrating power leaking in a time domainassociated with the channel frequency response of FIG. 8A. After a1024-point IFFT was performed, most of the energy is concentrated in atime domain window for the cyclic shift c=0, which is between timedomain indices 64 and 960 in FIG. 8B. However, there can be anon-negligible portion of power leaked into other windows.

Sidelobe leakage can degrade the channel frequency response of thecyclic shift that the sidelobe leakage is associated with, which can bedue to the loss of signal power in window truncating. Sidelobe leakagecan degrade the channel frequency response performance of one or moreother cyclic shifts by introducing inter-cyclic-shift interference.Therefore, ensuring that the channel impulse response of each cyclicshift is concentrated in its own time domain window can be significant.

Frequency offset will now be discussed. Even though a UE can remove mostof the frequency offset between its local oscillator and a localoscillator in a base station through an initial acquisition andfrequency tracking loop, there may still exist a residual frequencyoffset Δf. Sometimes, the residual frequency offset can be as large asseveral hundred Hertz.

Unlike the demodulation of a Physical Uplink Control Channel (PUCCH) andPhysical Uplink Shared Channel (PUSCH) where the frequency offset can beestimated and compensated for through a demodulation reference signal(DMRS), it can be more channeling to mitigate the frequency offset inSRS CE, especially when considering that different UEs may havedifferent frequency offset values.

FIG. 9 is a graph that illustrates frequency domain pulses with a timingoffset, a frequency offset, and no timing or frequency offset. Due tonon-zero frequency offset, a sinc pulse in the frequency domain can beshifted, for example, as shown in FIG. 9. The shifted sinc pulse isrepresented by sinc(f−(Δf/Bscs)), where Bscs is the subcarrier spacing.In FIG. 9, there is a leakage tap at the location of each subcarrier,resulting in inter-carrier interference (ICI). For a given frequencyoffset Δf, the ICI should become less severe with larger subcarrierspacing values.

Each leakage tap of the distorted sinc pulse contributes to a weightedand shifted version of ZC sequence, which can be represented by Equation11.

$\begin{matrix}\begin{matrix}{{A_{s} \cdot {x_{q}\left( {m - s} \right)}},} & {{m = 0},1,\ldots\mspace{20mu},{N_{ZC}^{RS} - 1}}\end{matrix} & \left( {{Eq}.\mspace{14mu} 11} \right)\end{matrix}$

Given the properties of ZC sequences discussed above, after de-covering,the frequency domain signal is a weighted channel frequency responsewith phase ramping, which can be represented by Equation 12.

$\begin{matrix}\begin{matrix}{{A_{s} \cdot {H(m)} \cdot e^{{- j}\frac{2{nsqm}}{N_{ZC}^{RS}}}},} & {{m = 0},1,\ldots\mspace{14mu},{N_{ZC}^{RS} - 1}}\end{matrix} & \left( {{Eq}.\mspace{14mu} 12} \right)\end{matrix}$

In the time domain, each leakage tap can cause a time domain spur. Thepower of the time domain spur can be related to the tap magnitude |As|.The locations of the spurs in time domain can be derived from the slopeof the phase ramping in frequency domain. For the s-th harmonic spur,the time domain location can be predicted by the Equation 13. Thelocation is a function of cell ID in Equation 13.

$\begin{matrix}\begin{matrix}{{{z\left( {s,q} \right)} = {{Mod}{\;\;}\left( {{{round}\left( {\frac{s \cdot q}{N_{ZC}^{RS}}N_{FFT}} \right)},N_{FFT}} \right)}},} & {{s = 0},{\underset{\_}{+}1},{\underset{\_}{+}2},{\underset{\_}{+}3},}\end{matrix} & \left( {{Eq}.\mspace{14mu} 13} \right)\end{matrix}$

The time domain spurs can cause interference across different cyclicshifts. The time domain spurs can degrade the SRS CE performance of eachUE.

Timing offset will now be discussed. A base station can adjust theuplink timing of a UE through timing advance (TA). However, due todithering in TA, non-zero timing offset may still exist between a symbolboundary of the base station and that of the received signal from a UE.

The timing offset may make it possible that fewer time domain samplesare selected in an FFT. In the frequency domain, this can result in thesinc pulse of the OFDM being distorted. Such distortion is shown in FIG.9. If there are ΔT samples missing in the FFT window, then in thefrequency domain, the distorted sinc pulse can be represented byFunction 1. Function 1 implies that there is a leakage tap at thelocation of each subcarrier.

$\begin{matrix}{{sinc}\left( {f\left( {1 - \frac{\Delta\; T}{N_{FFT}}} \right)} \right)} & \left( {{Fn}.\mspace{14mu} 1} \right)\end{matrix}$

The degradation caused by timing offset can be two-fold. First, theremay be inter-symbol interference (ISI) in the time domain. Second, thedistorted sinc pulse can cause ICI leakage in the frequency domain.

Similar to the frequency offset, the timing offset can introduceharmonic spurs in a time domain channel impulse response. The locationsof harmonic spurs can be predicted with the Equation 13.

Techniques to Improve Reference Signal Channel Estimation

Techniques to improve the performance of reference signal channelestimation are disclosed. These techniques can make reference signalchannel estimates more robust to one or more impairments. The improvedreference signal channel estimation can compensate for one or more ofdistortion of low-frequency tones, time-domain channel impulse responseleakage, frequency offset, or timing offset. Any suitable combination ofthe techniques to improve reference signal channel estimation disclosedherein can be implemented together with each other.

An example method of reference signal channel estimation includesde-covering a reference signal in a frequency domain to generate ade-covered reference signal; estimating a tone of the de-coveredreference signal based on at least two other tones of the de-coveredreference signal to generate an estimated tone; and generating a channelestimate based on the estimated tone and further processing, wherein thechannel estimate is associated with a wireless communication channelbetween a first node and a second node.

One technique to reduce distortion is tone estimation. Tone estimationcan involve interpolation and/or extrapolation. Estimation can be basedon poly-phase decomposition in some instances. Estimation can be basedon least-squared estimation in certain applications. Estimated tones canbe used to improve reference signal channel estimation. Tone estimationcan be performed, for example, using the tone estimation circuit 63 ofFIGS. 6A and/or 6C.

Linear interpolation and extrapolation based on poly-phase decompositionwill now be discussed. Equation 8 shows that, in the frequency domain,the de-covered SRS contains a sum of channel frequency response withdifferent linear phase ramping. In an example discussed below, it isassumed that n_(SRS) ^(cs,max)=8. Any other suitable value for n_(SRS)^(cs,max) can be used. For example, the extension to n_(SRS)^(cs,max)=12 is similar.

With n_(SRS) ^(cs,max)=8 in Equation 8, the de-covered SRS can berepresented by Equation 14.

$\begin{matrix}{{y^{\prime}(n)} = {{{y(n)} \cdot {{\overset{\_}{r}}^{*}(n)}} = {{\sum\limits_{c = 0}^{7}{e^{{j2}\;{\pi \cdot {(\frac{c}{8})} \cdot n}} \cdot {H_{c}(n)}}} + {v^{\prime}(n)}}}} & \left( {{Eq}.\mspace{14mu} 14} \right)\end{matrix}$

The value of

  e? ?indicates text missing or illegible when filed

repeats every 8 samples. Therefore, y′(n) can be decomposed into 8sub-sequences as shown in Equation 15.

$\begin{matrix}{{{y^{\prime}\left( {{8m} + p} \right)} = {{\sum_{c = 0}^{7}{e^{j\; 2\;{\pi \cdot {(\frac{c}{\pi})} \cdot p}} \cdot {H_{c}\left( {{8m} + p} \right)}}} + {v^{\prime}\left( {{8m} + p} \right)}}},{p = 0},1,\ldots\mspace{14mu},7.} & \left( {{Eq}.\mspace{14mu} 15} \right)\end{matrix}$

For a specific p, the sub-sequence y′(8m+p) should be smooth given theassumption that Hc(n) is slow-changing in the frequency domain.Accordingly, after the SRS de-covering, the distorted DC tones andadjacent tones can be linearly interpolated with the poly-phasedecomposition of y′(n).

FIG. 10 shows an example of interpolation. In FIG. 10, the center 8de-covered SRS tones can be replaced by 8 new tones interpolated from 16tones with 8 tones on each side. Each interpolated tone can be theaverage of two tones which are 8 tones from their respective sides inthis example with n_(SRS) ^(cs,max)=8. Interpolated tones can be usedfor the center group of 8 de-covered SRS tones. In some otherapplications, the center 8 de-covered SRS tones can be modified based onthe 8 new estimated tones.

FIG. 11 shows an example of extrapolation. Similar processing tointerpolation can be extended to edge extrapolation. FIG. 11 illustrateshow an extrapolated tone can be derived from two tones whose distancesfrom the extrapolated tones are multiples of n_(SRS) ^(cs,max) (i.e.,multiples of 8 in the illustrated example).

Interpolation and extrapolation based on least-square estimation willnow be discussed. In certain applications, least square estimation canbe implemented in place of estimation based on poly-phase decomposition.According to some applications, a system can select between least squareestimation and estimation based on poly-phase decomposition. Theavailable tones in a local region can be processed together tointerpolate distorted low frequency tones and/or to extrapolate for edgetones. In an example discussed below, it is assumed that n_(SRS)^(cs,max)=8. Any other suitable value for n_(SRS) ^(cs,max) can be used.For example, the extension to n_(SRS) ^(cs,max)=12 is similar.

In a local frequency region, (n) can be represented by a linearpolynomial as shown in Equation 16.

Hc(n)=a _(c,1) n+a _(c,0)  (Eq. 16)

There can be 2 unknown coefficients per cyclic shift. Consequently, theSRS of all cyclic shifts can be approximated by Equation 17. In Equation17, Ωcs represents the set of active cyclic shifts.

$\begin{matrix}{{y(n)} = {{\sum\limits_{c \in \Omega_{cs}}{e^{j\frac{\pi}{8}{cn}}\left( {{a_{c,1}n} + a_{c,0}} \right)}} + v_{n}}} & \left( {{Eq}.\mspace{14mu} 17} \right)\end{matrix}$

A coefficient vector for all 8 cyclic shifts can be denoted asa=[a_(0,1), a_(0,0), a_(1,1), a_(1,0), . . . , a_(7,1), a_(7,0)]^(T). Aset of L frequency domain samples for interpolation can be denoted as{(n), n ∈ ΩY}, ΩY={n₀, n₁, . . . , n_(L-1)}. Then Equation 17 can berepresented as Equation 18. In Equation 18, ⊗ denotes Kronecker product.

y=A·a+v,  (Eq. 18)

$A = \begin{pmatrix}{\left\lbrack {1,e^{j\frac{\pi}{8}n_{0}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}n_{0}}} \right\rbrack \otimes \left\lbrack {n_{0},1} \right\rbrack} \\{\left\lbrack {1,e^{j\frac{\pi}{8}n_{1}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}n_{1}}} \right\rbrack \otimes \left\lbrack {n_{1},1} \right\rbrack} \\\vdots \\{\left\lbrack {1,e^{j\frac{\pi}{8}n_{L - 1}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}n_{L - 1}}} \right\rbrack \otimes \left\lbrack {n_{L - 1},1} \right\rbrack}\end{pmatrix}$

Equation 19 provides a least-square solution to Equation 18.

a=(A ^(H) A)⁻¹ A ^(H) y  (Eq. 19)

The indices of K tones to be interpolated can be denoted as a setΩ_(I)={m₀, m₁, . . . , m_(K-1)}. A B matrix can be defined as follows.

$B = \begin{pmatrix}{\left\lbrack {1,e^{j\frac{\pi}{8}m_{0}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}m_{0}}} \right\rbrack \otimes \left\lbrack {m_{0},1} \right\rbrack} \\{\left\lbrack {1,e^{j\frac{\pi}{8}m_{1}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}m_{1}}} \right\rbrack \otimes \left\lbrack {m_{1},1} \right\rbrack} \\\vdots \\{\left\lbrack {1,e^{j\frac{\pi}{8}m_{K - 1}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}m_{K - 1}}} \right\rbrack \otimes \left\lbrack {m_{K - 1},1} \right\rbrack}\end{pmatrix}$

The interpolated low-frequency tones can be represented by Equation 20.The interpolation matrix E of Equation 21 is of dimension K×L and can bepre-calculated.

y _(LF) =Ba=Ey  (Eq. 20)

E=B(A ^(H) A)⁻¹ A ^(H)  (Eq. 21)

The assumption that Hc(n) is slow-changing in the frequency domain maynot be valid. To make interpolation more accurate, we can approximateHc(n) in a local frequency region by a quadratic polynomial shown inEquation 22.

H _(c)(n)=a _(c,2) n ² +a _(c,1) n+a _(c,0)  (Eq.22)

There can be 3 unknown coefficients per cyclic shift. Consequently, theSRS of all cyclic shifts can be approximated by Equation 23. In Equation23, Ωcs represents the set of active cyclic shifts.

$\begin{matrix}{{y(n)} = {{\sum_{c \in \Omega_{cs}}{e^{j\frac{\pi}{8}{cn}}\left( {{a_{c,2}n^{2}} + {a_{c,1}n} + a_{c,0}} \right)}} + v_{n}}} & \left( {{Eq}.\mspace{14mu} 23} \right)\end{matrix}$

The same procedure to derive the solution in Equations 20 and 21 can beapplied again to the quadratic polynomial assumption, with the exceptionof the coefficient vector which is replaced by a=[a_(0,2), a_(0,1),a_(0,0), a_(1,2), a_(1,1), a_(1,0), . . . , a_(7,2), a_(7,1),a_(7,0)]^(T) and the matrix A and B being replaced by:

$A = \begin{pmatrix}{\left\lbrack {1,e^{j\frac{\pi}{8}n_{0}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}n_{0}}} \right\rbrack \otimes \left\lbrack {n_{0}^{2},n_{0},1} \right\rbrack} \\{\left\lbrack {1,e^{j\frac{\pi}{8}n_{1}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}n_{1}}} \right\rbrack \otimes \left\lbrack {n_{1}^{2},n_{1},1} \right\rbrack} \\\vdots \\{\left\lbrack {1,e^{j\frac{\pi}{8}n_{L - 1}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}n_{L - 1}}} \right\rbrack \otimes \left\lbrack {n_{L - 1}^{2},n_{L - 1},1} \right\rbrack}\end{pmatrix}$ $B = {\begin{pmatrix}{\left\lbrack {1,e^{j\frac{\pi}{8}m_{0}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}m_{0}}} \right\rbrack \otimes \left\lbrack {m_{0}^{2},m_{0},1} \right\rbrack} \\{\left\lbrack {1,e^{j\frac{\pi}{8}m_{1}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}m_{1}}} \right\rbrack \otimes \left\lbrack {m_{1}^{2},m_{1},1} \right\rbrack} \\\vdots \\{\left\lbrack {1,e^{j\frac{\pi}{8}m_{K - 1}},\ldots\mspace{14mu},e^{j\frac{7\pi}{8}m_{K - 1}}} \right\rbrack \otimes \left\lbrack {m_{K - 1}^{2},m_{K - 1},1} \right\rbrack}\end{pmatrix}.}$

Various modifications are possible to the above formulation. Forexample, if the cyclic shifts are not fully used, the dimensions of aand A can be reduced. As another example, to approximate (n), differentcyclic shifts may take different orders of polynomials as desired. Themethods shown above for linear and/or quadratic interpolation based onleast squared estimation can be extended to the extrapolation of edgetones. The only difference can be the definitions of Ωcs, Ω_(Y), andΩ_(I).

FIGS. 12A, 12B, and 12C show examples with DC distortion, interpolation,and edge extrapolation. FIG. 12A shows a received frequency domain SRSafter an FFT is performed with distorted low-frequency tones. FIG. 12Bshows an SRS after applying quadratic interpolation to 16 low-frequencytones. This graph shows reduced distortion at low frequency compared tothe graph in FIG. 12A. FIG. 12C shows an SRS after applying quadraticextrapolation to 16 edge tones on each side. This graph also showsreduced distortion at low frequency compared to the graph in FIG. 12A.

Frequency domain channel frequency response pulse shaping can reducepower leakage due to side lobes and inter-cyclic shift interference canbe reduced. Channel frequency response pulse shaping can be performed,for example, using the channel frequency response pulse shaping circuit65 of FIGS. 6B and/or 6C.

Pulse shaping can be applied to the frequency-domain de-covered symbols.Edge tones can be multiplied by a function that is smooth in frequencydomain. Then after performing an inverse FFT, the channel impulseresponse of each cyclic shift should be more concentrated in its owntime-domain window. Accordingly, power leakage due to side lobes can bereduced and the inter-cyclic-shift interference can be lowered.

One example of the pulse is a raised-cosine function. The raised-cosinepulse can be applied to both the edge tones and extended tones on eachside of the band. FIG. 13A shows window functions for a rectangle pulseand a raised cosine pulse. FIG. 13B shows time domain tap powersrelative to a center tap for the rectangle and raised cosine pulses,respectively. FIG. 13B shows that pulse shaping by the raised cosinepulse reduces tap power leakage compared to using a rectangle pulse.

FIG. 14A is a plot of an SRS with extrapolated edge tones beforefrequency domain pulse shaping. FIG. 14B is a plot of a frequency domainpulse-shaping function. In FIG. 14B, a raised-cosine function spans fromsub-carrier indices −16 to 56 and 519 to 591. FIG. 14C is a plot of theSRS that includes extrapolated edge tones after frequency domain pulseshaping is applied. FIG. 14C shows reduced distortion as a result offrequency domain pulse shaping.

After each channel impulse response is separated in the time domain, anFFT can be applied to obtain the channel frequency response of eachcyclic shift in frequency domain. Then the channel frequency response ofeach edge tone can be scaled accordingly to compensate for the impact offrequency domain pulse shaping. The channel impulse response scalingcircuit 70 of FIG. 6E can perform such scaling.

Another technique to improve reference signal channel estimation is timedomain channel impulse response scaling based on a power delay profile.The scaling can be applied per tap. The scaling can be minimum meansquared error (MMSE) scaling. With power delay profiles, an averagepower on each tap can be measured and the noise power can be estimated.Then per-tap MMSE scaling can be applied to suppress noise taps. Thepower delay profile and MMSE scaling and include power delay filtering,noise power estimation, and per-tap scaling.

FIG. 15 is a diagram that illustrates time domain channel impulseresponses for cyclic shifts after an inverse FFT has been performed.This diagram is for 8 cyclic shifts. FIG. 15 shows that taps for noisepower estimation can be between windows for different cyclic shifts.

FIG. 16 is a flow diagram of an example method 160 of time domainchannel impulse response scaling based on a power delay profile. Themethod 160 can be performed, for example, using the channel impulseresponse scaling circuit 70 of FIG. 6E. At block 162, tap powers of atime domain taps are filtered. An infinite impulse response filter canperform the filtering. Equation 24 can represent the filtering, where nis a time domain index and a is the time constant of the filter.

P _(n) ←(1−α)· P _(n) +α·|h _(n)|²  (Eq. 24)

Noise power of taps is estimated at block 164. Noise power is estimatedbased on filtered tap powers {Pn}, by taking the average of selectedsub-carriers. Selected sub-carriers can be the taps for noise powerestimation indicated in FIG. 15.

At block 166, per-tap scaling is applied. The per-tap scaling caninvolve thresholding. In thresholding, taps with Pn>a·Pnoise areselected as channel taps. All other taps can be set to a value to removenoise, such as a value of 0. Per-cap scaling can involve MMSE scaling.In MMSE scaling, the n-th tap h_(n) is scaled by a factor

$\frac{\overset{\_}{P_{n}} - P_{noise}}{\overset{\_}{P_{n}}}.$

Any other suitable per-tap scaling technique can be applied.

FIG. 17 is a schematic block diagram of a time domain processing circuitthat can perform time domain channel impulse response scaling based on apower delay profile. The illustrated time domain processing circuitincludes a time domain filter 170 having a plurality of taps, a filter172, a noise estimation circuit 174, and a tap scaling circuit 176. Thefilter 172 can be an infinite impulse response filter. The filter 172can filter tap power of time domain taps. The noise power estimationcircuit 174 is configured to estimate noise power for a sub-set of thetaps of the filter corresponding to sub-carriers outside time windowsfor channel impulse responses. The tap scaling circuit 176 is configuredto perform a per-tap scaling on at least a portion of the taps based onthe estimated noise power. The tap scaling circuit 176 can perform pertap scaling based on thresholding, MMSE scaling, any other suitablescaling operation, or any suitable combination thereof.

Time domain pulse shaping can be applied to improve reference signalchannel estimation. To make the SRS CE less subject to frequency offsetand timing offset, TD pulse shaping can be applied to OFDM samplesbefore taking an FFT for frequency domain symbols.

FIG. 18 is a schematic block diagram of processing circuitry 180 withtime domain pulse shaping. The processing circuitry 180 implements partof the signal processing for received reference signals for generating achannel estimate. As illustrated, the processing circuitry 180 includesa time domain pulse shaping circuit 182, the Fast Fourier Transformblock 41, and the reference signal extraction circuit 42.

The time domain pulse shaping circuit 182 applies pulse shaping beforereceived time domain samples are converted to the frequency domain bythe Fast Fourier Transform block 41. This can reduce frequency offsetand/or timing offset. The time domain pulse shaping circuit 182 can beimplemented with any suitable processing circuitry disclosed herein. Anoutput of the time domain pulse shaping circuit 182 can be coupled to aninput of any of the Fast Fourier Transform blocks disclosed herein, suchas the Fast Fourier Transform block 31 of FIG. 3 and/or the Fast FourierTransform block 41 of FIG. 4A. The FFT block 41 can provide a frequencydomain signal to the reference signal extraction circuit 42 with areduced time offset and/or frequency offset due to the time domain pulseshaping by the time domain pulse shaping circuit 182.

Without the time domain pulse shaping circuit 182, a pulse applied to aFast Fourier Transform block can be a rectangle pulse. The time domainpulse shaping circuit 182 can apply a raised cosine pulse, for example.The time domain pulse shaping circuit 182 can apply any other suitablepulse that causes time offset and/or frequency offset to be reduced.

FIG. 19A is a graph of a rectangle pulse and a raised cosine pulse inthe time domain. The raised cosine pulse in FIG. 19A has a roll-offfactor of 1. The raised cosine pulse of FIG. 19A is an example of apulse that can be generated by the pulse shaping circuit 182 of FIG. 18to reduce timing offset and/or frequency offset. In one exampleapplication, the Fast Fourier Transform block 41 can generate a2048-point FFT. The teachings herein can be similarly applied to othersuitable FFT sizes. The pulse in frequency domain after an FFT ispreformed can be represented by Equation 25, in which f is normalized tosub-carrier spacing (SCS).

$\begin{matrix}{{h(f)} = \left\{ \begin{matrix}{{\frac{\pi}{2}{sinc}\;\left( \frac{1}{2} \right)f} = {\pm \frac{1}{2}}} \\{{{2 \cdot {{sinc}\left( {f/2} \right)}}\frac{\cos\left( {\pi\;{f/2}} \right)}{1 - {4f^{2}}}},{0.{w.}}}\end{matrix} \right.} & \left( {{Eq}.\mspace{14mu} 25} \right)\end{matrix}$

FIG. 19B is a graph of a sinc pulse and the raised cosine pulse in thefrequency domain. Compared to the sinc pulse, the decay of side lobes ofthe raised cosine pulse is significantly faster in the frequency domain,as shown in FIG. 19B. Consequently, the ICI leakage can be significantlysmaller in the frequency domain for the raised cosine pulse.

For a raised cosine pulse with a roll-off factor of 1, the decay of sidelobes is significantly faster in the frequency domain and ICI leakage issignificantly smaller except for the first harmonic spurs. By applyingtime domain pulse shaping, there can be a relatively large leakage tothe adjacent comb, for example, due to the doubled width of the mainlobe in frequency domain. Accordingly, it can be advantageous to applytime domain pulse shaping in applications where there is no SRS or otherreference signal being allocated to an immediately adjacent comb.

Frequency rotation with an offset can be applied to improve referencesignal channel estimation. This can reduce interference among users.

If a base station maintains an estimate of frequency offset for each UE,the estimate can come from PUSCH and/or PUCCH demodulation. In addition,if the base station can identify the UE having the dominant received SRSpower, most likely from the previous SRS CE, then in the time domain,the received samples can be rotated with the frequency offset of thedominant UE. Applying frequency rotation with one offset can be modeledby Equation 26.

y _(t)′(n)=y _(t)(n)·e ^(−f2πΔfT) ^(s) ^(n)  (Eq. 26)

In some applications, a frequency offset can be determined as the centerof mass for all cyclic shifts based on the estimated frequency offsetsand SRS powers of all UEs.

FIG. 20 is a schematic block diagram of processing circuitry 200 withfrequency rotation. The processing circuitry 200 implements part of thesignal processing for received reference signals for generating achannel estimate. As illustrated, the processing circuitry 200 includesa frequency rotation circuit 202, the Fast Fourier Transform block 41,and the reference signal extraction circuit 42.

The frequency rotation circuit 202 can perform frequency rotation tocompensate for a frequency offset. The frequency offset can be from adominant UE. The frequency offset can be from a center of mass of allcyclic shifts. The frequency rotation circuit 202 can reduce frequencyoffset. The frequency rotation circuit 202 can include a mixer. Thefrequency rotation circuit 202 can be implemented with any suitableprocessing circuitry disclosed herein. An output of the frequencyrotation circuit 202 can be coupled to an input of any of the FastFourier Transform blocks disclosed herein, such as the Fast FourierTransform block 31 of FIG. 3 and/or the Fast Fourier Transform block 41of FIG. 4A. The FFT block 41in FIG. 20 can provide a frequency domainsignal to the reference signal extraction circuit 42 with a reducedfrequency offset due to the frequency rotation by the frequency rotationcircuit 202.

Spurs can be moved into unused time space to improve reference signalchannel estimation. When detecting channel impulse responses for cyclicshifts, there can be unused time domain space that does not impactchannel estimation. Spurs and/or other noise can be moved to such unusedtime domain space to improve channel estimation.

FIG. 21 is a flow diagram of a method 210 of detecting cyclic shifts ofa reference signal in which spurs are moved into an unused time space. Acell identifier (ID) can be selected at block 212. When group andsequence hopping is disabled in the SRS, since the location of harmonicspurs can be predicted as a function of cell ID, a proper cell ID can beselected to make the first harmonic spurs fall into regions that are notused by any of the cyclic shifts.

Based on the cell ID selected at block 212, spurs (e.g., first harmonicspurs) are moved into regions outside the time domain windows for cyclicshifts at block 214. This technique can advantageously be appliedtogether with time domain pulse shaping (e.g., with the time domainpulse shaping circuit 182 of FIG. 18), since first harmonic spurs can bedominant after time domain pulse shaping is applied and such spurs canbe moved outside of the time domain windows for cyclic shifts in themethod 210. Cyclic shifts can be detected at block 216. This cyclicshift detection can be performed without the harmonic spurs that areoutside of the time domain windows for the cyclic shifts impactingchannel estimation. Channel estimation can be performed based on thedetected cyclic shifts and further processing.

Spurs can be moved into unused time space by the time domain processingcircuit 34 of FIG. 3 and/or the time domain processing and channelimpulse response separation circuit 48 of FIG. 4B.

As an example, if there are 4 SRS cyclic shifts in one comb with cyclicshifts c=0, 1, 2 and 3, by selecting a cell ID denoted as nCID such thatnCIDmod 30=15, the first harmonic spurs in the time domain can be movedinto a region outside of the truncation windows for the four cyclicshifts.

FIG. 22 illustrates the locations of the first harmonic spurs for fourcyclic shifts. As shown in FIG. 22, the harmonic spurs are outside ofthe time domain windows for the 4 cyclic shifts. In FIG. 22, the spursare moved into time domain indices below the time domain indices forwindows for the cyclic shifts.

As another example, if there are 8 SRS cyclic shifts in one comb, and ifthe time domain window size for each cyclic shift is set to 64, then thecell ID can be set to 12 or 17 such that the first harmonic spurs arelocated outside of the time-domain windows for all cyclic shifts. Therecan be smaller time domain windows with more cyclic shifts. In certaininstances, spurs can be moved to time domain indices between windows forcyclic shifts.

MIMO Environment

FIG. 23 is a diagram illustrating an example multiple-inputmultiple-output (MIMO) network environment 230 in which channelestimation based on a reference signal can be performed. Various UEs canwirelessly communicate with a network system in the MIMO networkenvironment 230. Such wireless communications can achieve highthroughputs. Antennas of MIMO network environment 230 for wirelesslycommunicating with UEs can be distributed. Channel estimates forchannels between different nodes can be performed in the MIMO networkenvironment 230 based on reference signal estimation using any suitabletechniques disclosed herein.

Various standards and/or protocols may be implemented in the MIMOnetwork environment 230 to wirelessly communicate data between a basestation and a wireless communication device. Some wireless devices maycommunicate using an orthogonal frequency-division multiplexing (OFDM)digital modulation scheme via a physical layer. Example standards andprotocols for wireless communication in the environment 230 can includethe third generation partnership project (3GPP) Long Term Evolution(LTE), Long Term Evolution Advanced (LTE Advanced), 3GPP New Radio (NR)also known as 5G, Global System for Mobile Communications (GSM),Enhanced Data Rates for GSM Evolution (EDGE), Worldwide Interoperabilityfor Microwave Access (WiMAX), and the IEEE 802.11 standard, which may beknown as Wi-Fi. In some systems, a radio access network (RAN) mayinclude one or more base stations associated with one or more evolvedNode Bs (also commonly denoted as enhanced Node Bs, eNodeBs, or eNBs),gNBs, or any other suitable Node Bs (xNBs). In some other embodiments,radio network controllers (RNCs) may be provided as the base stations. Abase station provides a bridge between the wireless network and a corenetwork such as the Internet. The base station may be included tofacilitate exchange of data for the wireless communication devices ofthe wireless network. A base station can perform reference signalchannel estimation is accordance with any suitable principles andadvantages disclosed herein.

A wireless communication device may be referred to as a user equipment(UE). The UE may be a device used by a user such as a smartphone, alaptop, a tablet computer, cellular telephone, a wearable computingdevice such as smart glasses or a smart watch or an ear piece, one ormore networked appliances (e.g., consumer networked appliances orindustrial plant equipment), an industrial robot with connectivity, or avehicle. In some implementations, the UE may include a sensor or othernetworked device configured to collect data and wirelessly provide thedata to a device (e.g., server) connected to a core network such as theInternet. Such devices may be referred to as Internet of Things (IoT)devices. A downlink (DL) transmission generally refers to acommunication from the base transceiver station (BTS) or eNodeB to a UE.An uplink (UL) transmission generally refers to a communication from theUE to the BTS.

FIG. 23 illustrates a cooperative, or cloud radio access network (C-RAN)environment 100. In the environment 230, the eNodeB functionality issubdivided between a base band unit (BBU) 240 and multiple remote radiounits (RRUs) (e.g., RRU 255, RRU 265, and RRU 275). The network systemof FIG. 23 includes the BBU 240 and the RRUs 255, 265, and 275. An RRUmay include multiple antennas, and one or more of the antennas may serveas a transmit-receive point (TRP). The RRU and/or a TRP may be referredto as a serving node. The BBU 240 may be physically connected to theRRUs such as via an optical fiber connection. The BBU 240 may provideoperational information to an RRU to control transmission and receptionof signals from the RRU along with control data and payload data totransmit. The RRU may provide data received from UEs within a servicearea associated with the RRU to the network. As shown in FIG. 23, theRRU 255 provides service to devices within a service area 250. The RRU265 provides service to devices within a service area 260. The RRU 275provides service to devices within a service area 270. For example,wireless downlink transmission service may be provided to the servicearea 270 to communicate data to one or more devices within the servicearea 270.

In the environment 230, a network system can wirelessly communicate withUEs via distributed MIMO. For example, the UE 283 can wirelesslycommunicate MIMO data with antennas of the network system that includeat least one antenna of the RRU 255, at least one antenna of the RRU265, and at least one antenna of the RRU 275. As another example, the UE282 can wirelessly communicate MIMO data with distributed antennas thatinclude at least one antenna of the RRU 255 and at least one antenna ofthe RRU 265. As one more example, the UE 288 can wirelessly communicateMIMO data with distributed antennas that include at least one antenna ofthe RRU 255 and at least one antenna of the RRU 275. Any suitableprinciples and advantages of the reference signal channel estimationdisclosed herein can be implemented in such distributed MIMOapplications, for example.

The illustrated RRUs 255, 265, and 275 include multiple antennas and canprovide MIMO communications. For example, an RRU may be equipped withvarious numbers of transmit antennas (e.g., 2, 4, 8, or more) that canbe used simultaneously for transmission to one or more receivers, suchas a UE. Receiving devices may include more than one receive antenna(e.g., 2, 4, etc.). An array of receive antennas may be configured tosimultaneously receive transmissions from the RRU. Each antenna includedin an RRU may be individually configured to transmit and/or receiveaccording to a specific time, frequency, power, and directionconfiguration. Similarly, each antenna included in a UE may beindividually configured to transmit and/or receive according to aspecific time, frequency, power, and direction configuration. Theconfiguration may be provided by the BBU 240.

The service areas shown in FIG. 23 may provide communication services toa heterogeneous population of user equipment. For example, the servicearea 250 may include a cluster of UEs 290 such as a group of devicesassociated with users attending a large event. The service area 250 canalso include an additional UE 292 that is located away from the clusterof UEs 290. A mobile user equipment 294 may move from the service area260 to the service area 270. Another example of a mobile user equipmentis a vehicle 186 which may include a transceiver for wirelesscommunications for real-time navigation, on-board data services (e.g.,streaming video or audio), or other data applications. The environment230 may include semi-mobile or stationary UEs, such as robotic device288 (e.g., robotic arm, an autonomous drive unit, or other industrial orcommercial robot) or a television 284, configured for wirelesscommunications.

A user equipment 282 may be located with an area with overlappingservice (e.g., the service area 250 and the service area 260). Eachdevice in the environment 230 may have different performance needs whichmay, in some instances, conflict with the needs of other devices.

Channel estimation in the network environment 230, such as estimation ofchannels between UEs and RRUs using reference signals in accordance withany suitable principles and advantages disclosed herein, can be robustto one or more of a variety of impairments. An accurate estimate for awireless communication channel can be useful for calibration and/or forprecoding.

Conclusion

Depending on the embodiment, certain acts, events, or functions of anyof the processes or algorithms described herein can be performed in adifferent sequence, can be added, merged, or left out altogether (e.g.,not all described operations or events are necessary for the practice ofthe process or algorithm). Moreover, in certain embodiments, operations,or events can be performed concurrently, e.g., through multi-threadedprocessing, interrupt processing, or multiple processors or processorcores or on other parallel architectures, rather than sequentially.

Conditional language used herein, such as, among others, “can,” “could,”“might,” “may,” “e.g.,” “such as,” and the like, unless specificallystated otherwise, or otherwise understood within the context as used, isgenerally intended to convey that certain embodiments include, whileother embodiments do not include, certain features, elements, and/oroperations. Thus, such conditional language is not generally intended toimply that features, elements, and/or operations are in any way requiredfor one or more embodiments or that one or more embodiments necessarilyinclude logic for deciding, with or without other input or prompting,whether these features, elements, and/or steps are included or are to beperformed in any particular embodiment. The terms “comprising,”“including,” and the like are synonymous and are used inclusively, in anopen-ended fashion, and do not exclude additional elements, features,acts, operations, and so forth. Additionally, the words “herein,”“above,” “below,” and words of similar import, when used in thisapplication, shall refer to this application as a whole and not to anyparticular portions of this application. Where the context permits,words in the above Detailed Description of Certain Embodiments using thesingular or plural may also include the plural or singular,respectively. Also, the term “or” is used in its inclusive sense (andnot in its exclusive sense) so that when used, for example, to connect alist of elements, the term “or” means one, some, or all of the elementsin the list.

Disjunctive language such as the phrase “at least one of X, Y, Z,”unless specifically stated otherwise, is otherwise understood with thecontext as used in general to present that an item, term, etc., may beeither X, Y, or Z, or any combination thereof (e.g., X, Y, and/or Z).Thus, such disjunctive language is not generally intended to, and shouldnot, imply that certain embodiments require at least one of X, at leastone of Y, or at least one of Z to each be present.

Unless otherwise explicitly stated or generally understood from context,articles such as “a” or “an” should generally be interpreted to includeone or more described items. Accordingly, phrases such as “a deviceconfigured to” are intended to include one or more recited devices. Suchone or more recited devices can also be collectively configured to carryout the stated recitations. For example, “a processor configured tocarry out recitations A, B and C” can include a first processorconfigured to carry out recitation A working in conjunction with asecond processor configured to carry out recitations B and C.

The word “coupled,” as generally used herein, refers to two or moreelements that may be either directly coupled to each other, or coupledby way of one or more intermediate elements. Likewise, the word“connected,” as generally used herein, refers to two or more elementsthat may be either directly connected, or connected by way of one ormore intermediate elements. Connections can be via an air interfaceand/or via wires and/or via optical fiber and/or via any other suitableconnection.

As used herein, the terms “determine” or “determining” encompass a widevariety of actions. For example, “determining” may include calculating,computing, processing, deriving, generating, obtaining, looking up(e.g., looking up in a table, a database or another data structure),ascertaining and the like via a hardware element without userintervention. Also, “determining” may include receiving (e.g., receivinginformation), accessing (e.g., accessing data in a memory) and the likevia a hardware element without user intervention. Also, “determining”may include resolving, selecting, choosing, establishing, and the likevia a hardware element without user intervention.

While the above detailed description has shown, described, and pointedout novel features as applied to various embodiments, it can beunderstood that various omissions, substitutions, and changes in theform and details of the devices or algorithms illustrated can be madewithout departing from the spirit of the disclosure. For example,circuit blocks and/or method blocks described herein may be deleted,moved, added, subdivided, combined, arranged in a different order,and/or modified. Each of these blocks may be implemented in a variety ofdifferent ways. Any portion of any of the methods disclosed herein canbe performed in association with specific instructions stored on anon-transitory computer readable storage medium being executed by one ormore processors. As can be recognized, certain embodiments describedherein can be embodied within a form that does not provide all of thefeatures and benefits set forth herein, as some features can be used orpracticed separately from others. The scope of certain embodimentsdisclosed herein is indicated by the appended claims rather than by theforegoing description. All changes which come within the meaning andrange of equivalency of the claims are to be embraced within theirscope.

What is claimed is:
 1. A method of reference signal channel estimation,the method comprising: receiving a reference signal for channelestimation; de-covering the reference signal in a frequency domain togenerate a de-covered reference signal; after the de-covering, frequencydomain processing the de-covered reference signal to cause distortion ofa direct current offset in the de-covered reference signal to bereduced; after the frequency domain processing, time domain processingto cause a noise floor associated with the de-covered reference signalto be reduced; and generating a channel estimate based on the frequencydomain processing and the time domain processing, wherein the channelestimate is associated with a communication channel between a first nodeand a second node.
 2. The method of claim 1, wherein the frequencydomain processing comprises generating an estimated tone for a tone ofthe de-covered frequency domain reference signal based on at least twoother tones of the de-covered frequency domain reference signal.
 3. Themethod of claim 2, wherein the frequency domain processing comprisesreplacing the tone with the estimated tone to cause distortionassociated with the direct current offset to be reduced.
 4. The methodof claim 2, wherein the frequency domain processing comprises modifyingthe tone based on the estimated tone to cause distortion associated withthe direct current offset to be reduced.
 5. The method of claim 1,wherein the frequency domain processing comprises pulse shaping thede-covered frequency domain reference signal to cause distortionassociated with the direct current offset to be reduced.
 6. The methodof claim 1, wherein the time domain processing comprises: estimatingnoise power for a sub-set of time domain taps corresponding tosub-carriers between channel impulse responses; and performing a per-tapscaling on at least a portion of the time domain taps based on theestimating.
 7. The method of claim 6, wherein the per-tap scalinginvolves at least one of minimum mean squared error scaling orthresholding.
 8. The method of claim 1, further comprising pulse shapingthe reference signal prior to the de-covering.
 9. The method of claim 1,further comprising rotating the reference signal based on an indicatorof a frequency offset prior to the de-covering.
 10. The method of claim1, wherein the time domain processing comprises moving a spur outside oftime domain windows for cyclic shifts of the reference signal.
 11. Themethod of claim 10, further comprising pulse shaping the referencesignal prior to the de-covering.
 12. The method of claim 1, wherein thereference signal is an uplink Sounding Reference Signal.
 13. The methodof claim 1, wherein the first node is a user equipment and the secondnode is a network node.
 14. A system for channel estimation, the systemcomprising: a frequency domain processing circuit configured to generatea de-covered frequency domain reference signal and to process thede-covered frequency domain reference signal so as to cause distortionassociated with a direct current offset to be reduced; and a time domainprocessing circuit having an input coupled to an output of the frequencydomain processing circuit, the time domain processing circuit configuredto suppress time domain channel impulse response leakage; and a channelestimation circuit configured to generate a channel estimate based on anoutput of the time domain processing circuit, wherein the channelestimate is associated with a wireless communication channel between afirst node and a second node.
 15. The system of claim 14, wherein thefrequency domain processing circuit is configured to generate anestimated tone for a tone of the de-covered frequency domain referencesignal based on at least two other tones of the de-covered frequencydomain reference signal, and to replace the tone with the estimated toneto cause distortion associated with the direct current offset to bereduced.
 16. The system of claim 14, wherein the frequency domainprocessing circuit is configured to perform pulse shaping on thede-covered frequency domain reference signal to cause distortionassociated with the direct current offset to be reduced.
 17. The systemof claim 14, wherein the time domain processing circuit comprises: afilter comprising a plurality of taps; a noise power estimation circuitconfigured to estimate noise power for a sub-set of the taps of thefilter corresponding to sub-carriers between channel impulse responses;and a filter tap scaling circuit configured to perform a per-tap scalingon at least a portion of the taps of the filter based on the estimatednoise power.
 18. The system of claim 14, wherein the time domainprocessing circuit is configured to move a spur outside of time domainwindows for cyclic shifts of the reference signal.
 19. The system ofclaim 14, further comprising a second time domain processing circuitconfigured to pulse shape a reference signal, the second time domainprocessing circuit having an output coupled to an input of the frequencydomain processing circuit.
 20. The system of claim 14, furthercomprising a second time domain processing circuit configured to rotatea reference signal based on an indicator of a frequency domain offset tothereby reduce the frequency domain offset, the second time domainprocessing circuit having an output coupled to an input of the frequencydomain processing circuit.
 21. The system of claim 14, furthercomprising a second frequency domain processing circuit configured toperform per-cyclic shift frequency domain processing, wherein the secondfrequency domain processing circuit is coupled between the time domainprocessing circuit and the channel estimation circuit.
 22. The system ofclaim 14, wherein the first node is a user equipment and the second nodeis a network node.
 23. A system for channel estimation, the systemcomprising: means for processing a de-covered frequency domain referencesignal so as to cause distortion associated with a direct current offsetto be reduced; means for suppressing time domain channel impulseresponse leakage, the means for suppressing having an input coupled toan output of the means for processing; and a channel estimation circuitconfigured to generate a channel estimate based on an output of themeans for suppressing, wherein the channel estimate is associated with awireless communication channel between a first node and a second node.